Digital transmitter

ABSTRACT

An equalizer provided in a digital transmitter compensates for attenuation in a signal channel to a digital receiver. The equalizer generates signal levels as a logical function of bit history to emphasize transition signal levels relative to repeated signal levels. The preferred equalizer includes an FIR transition filter using a look-up table. Parallel circuits including FIR filters and digital-to-analog converters provide a high speed equalizer with lower speed circuitry. The equalizer is particularly suited to in-cabinet and local area network transmissions where feedback circuitry facilitates adaptive training of the equalizer.

RELATED APPLICATIONS

This application is a continuation of application Ser. No. 11/483,971,filed Jul. 10, 2006, now abandoned which is a continuation ofapplication Ser. No. 10/372,630, filed on Feb. 24, 2003, now U.S. Pat.No. 7,099,404, which is a continuation of application Ser. No.09/852,481, filed on May 10, 2001, now U.S. Pat. No. 6,542,555, which isa continuation of Ser. No. 08/882,252, filed on Jun. 25, 1997, now U.S.Pat. No. 6,266,379, which is a continuation-in-part of Ser. No.08/880,980, filed on Jun. 23, 1997, now abandoned which claims thebenefit of U.S. Provisional Application No. 60/050,098, filed on Jun.20, 1997. The entire teachings of the above applications areincorporated herein by reference.

GOVERNMENT SUPPORT

The invention was supported, in whole or in part, by a grant No.F19628-92-C-0045 from Department of the Air Force. The Government hascertain rights in the invention.

BACKGROUND OF THE INVENTION

The performance of many digital systems is limited by theinterconnection bandwidth between chips, boards, and cabinets. As VLSItechnology continues to scale, system bandwidth will become an even moresignificant bottleneck as the number of I/Os scales more slowly than thebandwidth demands of on-chip logic. Also, off-chip signaling rates havehistorically scaled more slowly than on-chip clock rates. Most digitalsystems today use full-swing unterminated signaling methods that areunsuited for data rates over 100 MHz on one meter wires. Even goodcurrent-mode signaling methods with matched terminations and carefullycontrolled line and connector impedance are limited to about 1 GHz bythe frequency-dependent attenuation of copper lines. Without newapproaches to high-speed signaling, bandwidth will stop scaling withtechnology when we reach these limits.

SUMMARY OF THE INVENTION

Conventional approaches to dealing with frequency dependent attenuationon transmission lines have been based on equalization, either in thetransmitter or the receiver. For example, Tomlinson precoding is used inmodems, and digital equalization in binary communication channels hasbeen suggested in U.S. Pat. No. 4,374,426 to Burlage et al. However,such systems cannot scale to very high data rate binary or multilevelsystems having bandwidths extending from near DC to greater than 100MHz. Above 100 MHz, there is substantial attenuation due to skin effectresistance on conventional transmission lines.

The present invention enables equalizers which can be implemented asdigital filters operating at acceptable clock speeds. For example, athree gigabit per second (Gbps) system can be implemented using 400 Mbpscircuitry. The invention has particular application to nonmodulated,high data rate, binary or multilevel systems as found locally within adata processor cabinet or on a local area network.

In accordance with the present invention, a digital transmittercomprises an equalizer which emphasizes transition signal levelsrelative to repeated signal levels. In particular, a novel equalizergenerates signal levels as a logical function of bit history toemphasize transition signal levels. Preferred implementations define thelogical function of bit history in a look up table.

In preferred embodiments, the equalizer converts an input signal, havingdiscrete signal levels at an input data rate, to an output signal havinga greater number of discrete signal levels at the input data rate. Inparticular, the equalizer generates transmitted signal levels based ontime since last signal transition. A particularly simple implementationis based on whether a current bit is equal to an immediately previousbit.

The clock rates of circuitry can be reduced by multiplexing outputs ofparallel logic circuits operating on different multiple bit inputs togenerate the signal levels. In an adaptive system, the level ofequalization in the transmitter can be modified as a function of signalsdetected at the receiver.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing and other objects, features and advantages of theinvention will be apparent from the following more particulardescription of preferred embodiments of the invention, as illustrated inthe accompanying drawings in which like reference characters refer tothe same parts throughout the different views. The drawings are notnecessarily to scale, emphasis instead being placed upon illustratingthe principles of the invention.

FIG. 1 illustrates a digital communication system embodying in thepresent invention.

FIGS. 2A and 2B illustrate a sample binary pulse train and the resultantfrequency dependent attenuation caused by a transmission line.

FIGS. 3A and 3B illustrate the resistance and attenuation curves for onemeter of 30 AWG, 100 ohm twisted pair transmission line, and FIGS. 3Cand 3D illustrate the resistance and attenuation curves for one meter of5 mil 0.5 oz 50 ohm strip guide.

FIG. 4A illustrates respective plus and minus signals in a differentialsystem and the reduced data eye due to attenuation; FIG. 4B illustratestrailing edge jitter; and FIG. 4C illustrates the data eye withequalization.

FIGS. 5A and 5B illustrate impulse response and frequency response of anequalizing filter embodying the invention, and FIGS. 5C and 5Dillustrate an example input sequence and output sequence from theequalizer.

FIG. 6A illustrates the frequency response of an equalization filterembodying the invention; FIG. 6B illustrates transmission lineattenuation; and FIG. 6C illustrates the combination of equalization andline attenuation.

FIG. 7A illustrates an equalized transmitter signal based on the inputsignal of FIG. 2A, and FIG. 7B illustrates the signal at the receiverresulting from the signal of FIG. 7A to be compared to FIG. 2B withoutequalization.

FIG. 8 illustrates one embodiment of an equalizer of the presentinvention including an FIR filter and digital-to-analog converter.

FIG. 9 illustrates a transition filter for use in a preferred embodimentof the invention.

FIG. 10 illustrates a two tap transition filter embodying the invention.

FIGS. 11A and 11B illustrate a digital to analog converter for use inthe present invention.

FIG. 12 illustrates a preferred multiplexed embodiment of the invention.

FIG. 13 illustrates CPU to memory and peripheral connections embodyingthe present invention.

DETAILED DESCRIPTION OF THE INVENTION

A description of preferred embodiments of the invention follows.

The density and speed of modern VLSI technology can be applied toovercome the I/O bottleneck they have created by building sophisticatedI/O circuitry that compensates for the characteristics of the physicalinterconnect and cancels dominant sources of timing and voltage noise.Such optimized I/O circuitry is capable of achieving I/O rates an orderof magnitude higher than those commonly used today while operating atlower power levels.

A system embodying the invention can achieve a four Gbps signaling rateusing 0.5 μm CMOS circuits by controlling and compensating forcharacteristics of the transmission medium, by cancelling timing skew,and through careful management of time and voltage noise.

FIG. 1 shows one channel of high-speed signaling system embodying theinvention. A transmitter module 22 accepts 8-bit parallel data at 400MHz. Each byte is coded into 10 bits for band-limiting and forward errorcorrection and transmitted up to 3 m across a single differentialtransmission line. The transmitter pre-emphasizes the signal tocompensate for expected line characteristics. The lossy transmissionline as well as package and connector parasitics attenuate and distortthe received waveform, and it is further corrupted by noise coupled fromadjacent lines and the power supply. The receiver 24 accepts this noisy,distorted signal and its own 400 MHz clock. The receiver generates 4 GHztiming signals aligned to the received data, samples the noisy signal,decodes the signal, and produces synchronous 8-bit data out.

The availability of 4 Gbps electrical signaling will enable the designof low-cost, high-bandwidth digital systems. The wide, slow buses aroundwhich many contemporary digital systems are organized can be replaced bypoint-to-point networks using a single, or at most a few, high-speedserial channels resulting in significant reduction in chip and modulepinouts and in power dissipation. A network based on 400 MBytes/s serialchannels, for example, has several times the bandwidth of a 133 MBytes/sPCI-bus that requires about 80 lines. Also, depending on its topology,the network permits several simultaneous transfers to take place at fullrate. A group of eight parallel channels would provide sufficientbandwidth (3.2 GBytes/s) for the CPU to memory connection (FIG. 13) oftoday's fastest processors. For modest distances (up to 30 m with 18 AWGwire), high-speed electrical signaling is an attractive alternative tooptical communication in terms of cost, power, and board area forperipheral connection (FIG. 13) and building-sized local-area networks.

Frequency-Dependent Attenuation Causes Intersymbol Interference

Skin-effect resistance causes the attenuation of a conventionaltransmission line to increase with frequency. With a broadband signal,as typically used in digital systems, the superposition of unattenuatedlow-frequency signal components with attenuated high-frequency signalcomponents causes intersymbol interference that degrades noise marginsand reduces the maximum frequency at which the system can operate.

This effect is most pronounced in the case of a single 1 (0) in a fieldof 0s (1s) as illustrated in FIGS. 2A and B. The figures show a 4 Gb/ssignal (FIG. 2A) and the simulated result of passing this signal across3 m of 24 AWG twisted pair (FIG. 2B). The highest frequency of interest(2 GHz) is attenuated by −7.6 dB (42%). The unattenuated low-frequencycomponent of the signal causes the isolated high-frequency pulse tobarely reach the midpoint of the signal swing giving no eye opening in adifferential system and very little probability of correct detection.

The problem here is not the magnitude of the attenuation, but rather theinterference caused by the frequency-dependent nature of theattenuation. The high-frequency pulse has sufficient amplitude at thereceiver for proper detection. It is the offset of the pulse from thereceiver threshold by low-frequency interference that causes theproblem. Later, we will see how using a transmitter equalizer topreemphasize the high-frequency components of the signal eliminates thisproblem. However, first we will characterize the nature of thisattenuation in more detail.

FIGS. 3A-D show the resistance per meter and the attenuation per meteras a function of frequency for a 30 AWG (d=128 mm) twisted pair with adifferential impedance of 100 ohms (FIGS. 3A and 3B) and for a 5 mil(d=125 mm) half-ounce (0.7 mil thick) 50 ohms (FIGS. 3C and 3D)stripguide. For the 30 AWG pair, the skin effect begins increasingresistance at 267 KHz and results in an attenuation to 56% of theoriginal magnitude (−5 dB) per meter of cable at our operating frequencyof 2 GHz corresponding to a bit rate of 4 Gb/s. Skin effect does notbegin to effect the 5 mil PC trace until 43 MHz because of its thinvertical dimension. The high DC resistance (6.8 ohms/m) of this linegives it a DC attenuation of 88% (−1.2 dB). Above 70 MHz the attenuationrolls off rapidly reaching 40% (−8 dB) at 2 GHz. The importantparameter, however, is the difference between the DC and high-frequencyattenuation which is 45% (−6.8 dB).

The effect of frequency dependent attenuation is graphically illustratedin the eye-diagrams of FIG. 4A-C. As shown in the waveform in FIG. 4A,without equalization, a high-frequency attenuation factor of A reducesthe height of the eye opening to 2A-1 with the eye completelydisappearing at A≦0.5. This height is the amount of effective signalswing available to tolerate other noise sources such as receiver offset,receiver sensitivity, crosstalk, reflections of previous bits, andcoupled supply noise. Because the waveforms cross the receiver thresholdoffset from the center of the signal swing, the width of the eye is alsoreduced. As illustrated in FIG. 4B, the leading edge of the attenuatedpulse crosses the threshold at the normal time. The trailing edge,however, is advanced by t_(j). This data-dependent jitter causes greatersensitivity to skew and jitter in the signal or sampling clock and mayintroduce noise into the timing loop.

The waveform of FIG. 4C illustrates the situation when we equalize thesignal by attenuating the DC and low frequency components so allcomponents are attenuated by a factor of A. Here the height of the eyeopening is A, considerably larger than 2A-1, especially for largeattenuations. Also, because the waveforms cross at the midpoint of theirswing, the width of the eye is a full bit-cell giving better toleranceof timing skew and jitter.

Preemphasizing Signal Transitions Equalizes Line Attenuation

Equalization eliminates the problem of frequency-dependent attenuationby filtering the transmitted or received waveform so the concatenationof the equalizing filter and the transmission line gives a flatfrequency response. With equalization, an isolated 1 (0) in a field of0s (1s) crosses the receiver threshold at the midpoint of its swing, asshown in FIG. 4C, rather than being offset by an unattenuated DCcomponent, as shown in FIG. 4A. Narrow-band voice, video, and datamodems have long used equalization to compensate for the linear portionof the line characteristics (Lee, Edward A., and Messerschmitt, DavidG., Digital Communication, Second Edition, Kluwer, 1994). However, ithas not been used to date in broadband signaling with a wide bandwidth(i.e., greater than 100 MHz) over short distances.

We equalize the line using a 4 GHz FIR filter built into thecurrent-mode transmitter. The arrangement is similar to the use ofTomlinson precoding in a narrowband modem (Tomlinson, M., “New AutomaticEqualizer Employing Modulo Arithmetic,” Electronic Letters, March 1971).In a high-speed digital system it is much simpler to equalize at thetransmitter than at the receiver, as is more commonly done incommunication systems. Equalizing at the transmitter allows us to use asimple receiver that just samples a binary value at 4 GHz. Equalizing atthe receiver would require an A/D of at least a few bits resolution or ahigh-speed analog delay line, both difficult circuit design problems. Adiscrete-time FIR equalizer is preferable to a continuous-time passiveor active filter as it is more easily realized in a standard CMOSprocess.

After much experimentation we have selected a five-tap FIR filter thatoperates at the bit rate. The weights are trained to match the filter tothe frequency response of the line as described below. For a 1 m 30 AWGline, the impulse response is shown in FIG. 5A. Each vertical linedelimits a time interval of one bit-cell or 250 ps. The filter has ahigh-pass response as shown in FIG. 5B.

As shown in FIGS. 6A-C, this filter cancels the low-pass attenuation ofthe line giving a fairly flat response over the frequency band ofinterest (the decade from 200 MHz to 2 GHz). We band-limit thetransmitted signal via coding to eliminate frequencies below 200 MHz.The equalization band is limited by the length of the filter. Addingtaps to the filter would widen the band. We have selected five taps as acompromise between bandwidth and cost of equalization.

FIG. 6A shows the frequency response of the filter, FIG. 6B shows thefrequency response of the line and FIG. 6C shows the combination (theproduct) for 1 m of 30 AWG cable. The scale on FIG. 6C is compressed toexaggerate the effect. The filter cancels the response of parasitics aswell as the response of the line. The response is flat to within 5%across the band of interest. The filter results in all transitions beingfull-swing, while attenuating repeated bits. FIG. 5D shows the responseof the filter to an example data sequence shown in FIG. 5C(00001000001010111110000). The example shows that each signal transitiongoes full swing with the current stepped down to an attenuated level forrepeated strings of 1s (0s).

FIGS. 7A and B illustrate the application of equalization to the exampleof FIGS. 2A and 2B. FIG. 7A shows the filtered version of the originalsignal and FIG. 7B the received waveform. With equalization the isolatedpulses and high-frequency segments of the signal are centered on thereceiver threshold and have adequate eye openings for detection.

Circuit Implementations

Preferred implementations of the invention include finite input response(FIR) filters, and FIG. 8 illustrates one such implementation. In thiscase, a 5 tap filter has been selected as a balance between higherfractional bandwidth and circuit complexity. With a greater number oftaps, equalization can be obtained at lower frequencies. The presentdesign provides for equalization in a range of 100 MHz to 2 GHz. Byreducing to 2 or 3 taps, the lower end of the range may be no less than500 MHz.

As in a conventional FIR filter, the input D_(i) is delayed insuccessive delay elements 28. However, rather than weighting theindividual delayed signals and summing the weighted signals to obtainthe desired output, the delayed signals are applied to a 5-to-32 decoder32.

One of the 32 output bits from the decoder 32 is high with any inputstate and that high bit addresses a 4 bit word from the 32×4 randomaccess memory 34. The memory 34 is shown to be random access in order toallow for reprogramming of the equalization using a training processbelow. However, the system may be a fixed design which can beimplemented using a read only memory.

The 4 bit output from RAM 34 defines one of the 15 output levelsgenerated by a digital-to-analog converter 36 and applied to thetransmission line 38. Those levels include 0, seven positive levelswhere Dout− is pulled low, and seven negative levels where Dout+ ispulled low. To simplify the implementation, each FIR filter isapproximated by a transition filter implemented with a look-up table asillustrated in FIG. 9. The transition filter compares, in logic elements40, the current data bit D_(i) to each of the last four bits, and uses afind-first-one unit 42 to determine the number of bits since the lastsignal transition. The result is used to look up a 3-bit drive strengthfor the current bit from a 15-bit serially-loaded RAM 44. The drivestrength is multiplied by the current bit with two sets of three NANDgates 46, 48 to generate three-bit high and low drive signals for theDAC.

While the transition filter is a non-linear element, it closelyapproximates the response of an FIR filter for the impulse functionsneeded to equalize typical transmission lines. Making this approximationgreatly reduces the size and delay of the filter as a 96-bit RAM wouldbe required to implement a full 5-tap FIR filter via a lookup table andthe gates 46 and 48.

The transition filter can be simplified even further to the simple logiccircuit of FIG. 10 which operates as a two tap filter. The input signalD_(i) is delayed in a single delay element 50 to produce the signalD_(i−1). The two signals are combined in an exclusive-OR gate 52 todetermine whether the current bit is equal to the immediately previousbit. If so, the lower magnitude output is generated by thedigital-to-analog converter 54. If, on the other hand, there has been atransition since the previous bit, the output is emphasized. Thus, thissimple circuit provides four output levels, two positive and twonegative.

In yet another two-tap embodiment, with a transition, full current driveis used in opposite directions on both sides of the transition. When thesignal value remains unchanged, an attenuated current drive is used.

The circuit design of the DAC used in the FIG. 9 embodiment is shown inFIGS. 11A and B. As shown in FIG. 11A, each DAC module is composed ofthree progressively sized differential pulse generators 56, 58 and 60.Each generator is enabled to produce a current pulse on Dout+ (Dout−) ifthe corresponding H (L) line is low. If neither line is low no pulse isproduced. Depending on the current bit and the three-bit value read fromthe RAM 44 in the filter module, 15 different current values arepossible (nominally from −8.75 mA to +8.75 ma in 1.25 mA steps). Thetiming of the pulse is controlled by a pair of clocks. A low-goingon-clock φ_(i) gates the pulse on its falling edge. The high-true offclock φ_(i+1) gates the pulse off 250 ps later.

Each of the three differential pulse generators is implemented as shownin FIG. 11B. A pre-drive stage 62 inverts the on-clock in inverter 64and qualifies the off-clock with the enable signals in NOR gates 66 and68. A low (true) enable signal, which must be stable while the off-clockis low, turns on one of the two output transistors 70, 72, priming thecircuit for the arrival of the on-clock. When the on-clock falls, thecommon tail transistor 74 is turned on, starting the current pulse. Whenthe off-clock rises, the selected output transistor terminates thecurrent pulse. The delay of the qualifying NOR-gate is carefully matchedagainst that of the on-clock inverter to avoid distorting the pulsewidth.

To enable operation of the equalization circuit at rates in the order ofgigahertz while using circuitry operating only in the order of hundredsof megahertz, the preferred embodiment generates the signal levels bymultiplexing outputs of parallel logic circuits operating on differentmultiple bit inputs.

A block diagram of the multiplexed transmitter is shown in FIG. 12. Thetransmitter accepts 10 bits of data, D₀₋₉, at 400 MHz. A distributionblock 76 delivers 5 bits of data to each of 10 FIR filters 78. The ithfilter receives bit D_(i) and the four previous bits. For the first fourfilters this involves delaying bits from the previous clock cycle. Thedistribution also retimes the filter inputs to the clock domain of thefilter. Each filter 78 is a 5-tap transition filter that produces a4-bit output encoded as 3 bits of positive drive and 3 bits of negativedrive. These six bits from the filter directly select which of six pulsegenerators in the DAC 80 connected to that filter are enabled. Theenabled pulse generators are sequenced by the 10-phase clock 82. The ithpulse generator is gated on by φ_(i) and gated off by φ_(i+1). To meetthe timing requirements of the pulse generator, the ith filter operatesoff of clock φ_(i+1).

A training sequence may be used to initialize the transmitterpre-emphasis filter at powerup. Training is performed under the controlof a supervisory processor 26 that interfaces with the transmitter onone end of the line and the receiver on the other end via a low-speedserial scan chain. A preliminary version of a training sequence for onechannel is as follows:

-   -   1. The frequency response of the line is measured. The        transmitter is commanded to turn off precompensation and send an        alternating sequence of 1s and 0s. The receiver measures the        level of the received signal by using a feedback transmitter to        shift the DC operating point of the sense-amplifiers. The        process is repeated at other bit rates to trace out the        attenuation curve. For example, bit rates of R_(max), R_(max)/2,        R_(max)/3 . . . may be tested.    -   2. Based on the attenuation measurements taken in (1), the        transmitter equalization is set by programming the FIR filter        and/or DAC.

CONCLUSION

Transmitter equalization extends the data rates and distances over whichelectronic digital signaling can be reliably used. Preemphasizing thehigh-frequency components of the signal compensates for the low-passfrequency response of the package and transmission line. This preventsthe unattenuated low-frequency components from interfering withhigh-frequency pulses by causing offsets that prevent detection. Withequalization an isolated pulse at the receiver has the same amplitude asa long string of repeated bits. This gives a clean received signal witha good eye opening in both the time and voltage dimensions.

In one embodiment, we implement equalization for a 4 Gbs signalingsystem by building a 4 GHz, five-tap FIR filter into the transmitter.This filter is simple to implement yet equalizes the frequency responseto within 5% across the band of interest. The filter is realized using0.5 mm CMOS circuitry operating at 400 MHz using a bank of 10 filtersand DACs sequenced by a 10-phase 400 MHz clock. Narrow drive periods arerealized using series gating to combine two clock phases, an on-phaseand off-phase, in each DAC. We have simulated extracted layout of theequalized transmitter driving a load through package parasitics and 1 mof differential strip guide to demonstrate the feasibility of thisapproach.

The equalizing transmitter described here is one component of a 4 Gbssignaling system we are currently developing for implementation in an0.5 μm CMOS technology. The system also relies on low-jitter timingcircuitry, automatic per-line skew compensation, a narrow-aperturereceive amplifier, and careful package design.

The availability of 4 Gbs serial channels in a commodity CMOS technologywill enable a range of system opportunities. The ubiquitous system buscan be replaced by a lower-cost yet higher-speed point-to-point network.A single hub chip with 32 serial ports can directly provide theinterconnection for most systems and can be assembled into moresophisticated networks for larger systems. A single 4 Gbs serial channelprovides adequate bandwidth for most system components and multiplechannels can be ganged in parallel for higher bandwidths.

A 4 Gbs serial channel can also be used as a replacement technology atboth the component and system level. At the component level, a singleserial channel (two pins) replaces 40 100 MHz pins. A 4 GByte/s CPU toL2 cache interface, for example, could be implemented with just eightserial channels. At the system level, high-speed electrical serialchannels are a direct replacement for expensive optical interconnect.Using 18 AWG wire, these channels will operate up to lengths of 10 menabling high-bandwidth, low-cost peripheral connections and local-areanetworks. Inexpensive electrical repeaters can be used to operate oversubstantially longer distances.

Even with 4 Gbs channels, system bandwidth remains a major problem forsystem designers. On-chip logic bandwidth (gates×speed) is increasing ata rate of 90% per year (60% gates and 20% speed). The density andbandwidth of system interconnect is increasing at a much slower rate ofabout 20% per year as they are limited by mechanical factors that are ona slower growth curve than that of semiconductor lithography. A majorchallenge for designers is to use scarce system interconnect resourceseffectively, both through the design of sophisticated signaling systemsthat use all available wire bandwidth and through system architecturesthat exploit locality to reduce the demands on this bandwidth.

While this invention has been particularly shown and described withreferences to preferred embodiments thereof, it will be understood bythose skilled in the art that various changes in form and details may bemade therein without departing from the spirit and scope of theinvention as defined by the appended claims.

1. A circuit comprising: a semiconductor chip; a transmitter circuitdisposed on the chip, the transmitter circuit being operable to accept adigital input signal including a plurality of bits and send an outputsignal including a series of output bit signals, each bit of the digitalinput signal being represented by a single output bit signal, eachoutput bit signal having a signal level which is a first function of thevalue represented by such output bit signal and the values of the bitsrepresented by one or more preceding output bit signals, the transmittercircuit including an input signal circuit operable to accept bits of thedigital input signal, a memory operable to store a plurality of dataelements and to retrieve a particular data element responsive to anaddress which is a second function of a current bit and one or morepreceding bits of the digital input signal, and a signal generatorcircuit operable to set the signal level of the output bit signalrepresenting the current bit responsive to the retrieved data element.2. The circuit as claimed in claim 1 wherein the transmitter circuitfurther includes an address derivation circuit operable to derive anaddress from the current bit and one or more preceding bits and whereinthe memory is operable to retrieve the output data element responsive tothe address derived by the address derivation circuit.
 3. The circuit asclaimed in claim 1 wherein the first function is such that an output bitsignal representing a bit having a value different from the value of thebits represented by one or more preceding output bit signals has asignal level of greater magnitude than an output bit signal representinga bit having the same value as the bits represented by one or morepreceding output bit signals.
 4. The circuit as claimed in claim 1wherein the transmitter circuit is operable to provide the output signalwith an output frequency of at least 1 GHz and a bandwidth greater than100 MHz.
 5. The circuit as claimed in claim 1 wherein the first functionis fixed.
 6. The circuit as claimed in claim 5 wherein the memoryincludes a read write memory.
 7. The circuit as claimed in claim 1wherein the first function is reprogrammable.
 8. The circuit as claimedin claim 1 wherein the first function is adaptive.
 9. The circuit asclaimed in claim 1 wherein the memory includes a read write memory. 10.The circuit as claimed in claim 1 wherein the signal generator circuitincludes a digital to analog converter.
 11. The circuit as claimed inclaim 10 wherein the digital to analog converter selects signal levelsbased on only the retrieved data element.
 12. The circuit as claimed inclaim 10 wherein the digital to analog converter selects signal levelsresponsive to the retrieved data element and to the current bit.
 13. Thecircuit as claimed in claim 10 wherein the digital to analog converterincludes a plurality of current pulse drivers connected to a commonoutput, and wherein the transmitter circuit includes a circuit operableto select one or more of the current pulse drivers for each output bitsignal and actuate the selected ones of the current pulse drivers toprovide the output bit signal as a current pulse.
 14. The circuit asclaimed in claim 10 wherein the signal generator circuit is responsiveto the current bit to set a sign of the signal level and responsive tothe retrieved data element to set a magnitude of the signal level. 15.The circuit as claimed in claim 1 further comprising a second memory,the transmitter circuit being coupled to the second memory to accept thedigital input signal from the second memory.
 16. The circuit as claimedin claim 1 further comprising a processor, the transmitter circuit beingcoupled to the processor to accept the digital input signal from theprocessor.
 17. The circuit as claimed in claim 1 further comprising aperipheral, the transmitter circuit being coupled to the peripheraldevice to accept the digital input signal from the peripheral.
 18. Anapparatus comprising a plurality of elements selected from the groupconsisting of processors, additional memories, and peripherals, at leastone signal path for conveying information between the elements, and atleast one circuit as claimed in claim 1, the transmitter circuit beingcoupled to one of the elements for acceptance of the digital inputsignal from that one of the elements and being coupled to the signalpath for transmission of the output bit signals along the signal path toanother one of the elements.
 19. A method of sending informationcomprising the steps of: (a) accepting a digital input signal includinga plurality of bits at a transmitter circuit within a semiconductorchip; and (b) sending an output signal from the transmitter circuit, theoutput signal including a series of output bit signals, each bit of thedigital input signal being represented by a single output bit signal,each output bit signal having a signal level, the sending step includingthe steps of (i) deriving an address associated with each output bitsignal from a current bit of the input signal to be represented by thatoutput bit signal and one or more bits of the input signal representedby preceding output bit signals, (ii) responsive to the address,retrieving a particular data element associated with such output bitsignal from a memory storing a plurality of data elements, and (iii)setting the signal level of the output bit signal representing thecurrent bit responsive to the retrieved data element.
 20. The method asclaimed in claim 19 wherein an output bit signal representing a bithaving a value different from the value of the bits represented in oneor more preceding output bit signals has a signal level of greatermagnitude than an output bit signal representing a bit having the samevalue as the bits represented by one or more preceding output bitsignals.
 21. The method as claimed in claim 19 wherein the steps ofaccepting the digital input signal and sending the output signal conveyinformation along a signal path of a digital system between two or moreelements of the digital system selected from the group consisting ofprocessors, additional memories, hubs, and peripherals.
 22. The methodas claimed in claim 21 wherein the signal path has a length of up to 10meters.
 23. The method as claimed in claim 19 wherein the output signalhas a maximum frequency of at least 1 GHz and a bandwidth greater than100 MHz.
 24. The method as claimed in claim 19 wherein the memory isfixed.
 25. The method as claimed in claim 19 further comprising the stepof reprogramming the memory.
 26. The method as claimed in claim 19further comprising the steps of monitoring the output signal aftertransmission along a signal path, and reprogramming the memoryresponsive to the results of the monitoring step.